Converter circuit and control method for same

ABSTRACT

In known converter circuits switching losses occur, which are caused by reverse-recovery currents of a freewheeling diode. To reduce said switching losses it is proposed by the invention to drive the switching elements such that, upon switching from the second to the first switching element, the timing is controlled in such a manner that the shoot through currents and the conduction of the freewheeling diode are kept at a low value or, better still, are precluded. As regards the control mechanism, it is proposed to turn on the first switching element later if shoot through currents occur, and to turn on the first switching element sooner if conduction of the freewheeling diode occurs. Here, a time of overlap may be provided during which both switching elements are simultaneously conducting. For the control mechanism, the voltage across a switching element can be used as a measured input value.

The invention relates to a converter circuit and a control method aswell as to a drive device for a converter circuit.

Converter circuits are used to convert an input voltage to an outputvoltage. Particularly for DC/DC converter circuits a plurality oftopologies are known, i.e. circuits in which the switches employedtherein are adequately driven so as to fulfill various requirements.

A known converter topology is the synchronous step-down converter (buckconverter). On the side of its input, said buck converter comprises ahalf bridge with a first switching element (control) and a secondswitching element (sync), which half bridge is operated at an input DCvoltage. The bridge arm connected between the switching elementscomprises an inductor behind which the load is connected. The switchingelements are alternately driven, for example, by means of a pulseduration control. Buck converters are used for a wide range ofapplications, including VRMs (Voltage Regulator Modules) for modernmicroprocessors.

The switching elements customarily comprise freewheeling diodes. Whenfield effect transistors are used as switching elements, thefreewheeling diode is part of the switching element, i.e. the diodebetween drain and source. When half bridges are driven, dead times arecustomarily set between the switching operations of the switchingelements so as to preclude that conduction of the switching elements andthe resultant shoot through currents (shoot-through) occursimultaneously. After turn-off of a switching element there is always adead time in which both switching elements are turned off. In this deadtime the current, which necessarily must be a continuous current due tothe inductance on the output side, is maintained in that a freewheelingdiode is conducting. As regards switching between the switchingelements, a distinction can be made between “hard” and “soft” switchingtransitions (soft-switching). In the case of hard switching transitions,a switching element is turned on while a voltage is applied across theswitching element. In the case of soft switching transitions, alsoreferred to as ZVS (Zero Voltage Switching), no or only a very smallvoltage is applied across a switching element when said switchingelement is turned on. If the buck converter is operated in a manner suchthat the load current through the inductance does not change polarity(continuous operation), then a behavior is obtained such that a “hard”switching transition (switching from sync switch to control switch) anda ZVS transition (switching from control switch to sync switch) occurswithin each switching period.

To achieve a further miniaturization of converter circuits as well as avery fast reaction to load alternations, high switching frequencies aredesirable. What is problematic in this respect is that the switchinglosses increase with the frequency. A substantial part of theseswitching losses originates from the reverse current (reverse recoverycurrent) of the freewheeling diode of the synchronous switch. In thecase of synchronous buck converters, said reverse current accompanieshard switching transitions from the sync switch to the control switch.This problem is encountered also with other converter topologies.

In U.S. Pat. No. 5,539,630, known DC/DC switching converters areexamined in respect of their drawbacks. As regards the buck converter,the reverse recovery problem is mentioned, which should be addressed bymeans of a circuit with a magnetic saturation element.

Also U.S. Pat. No. 5,479,089 considers various converter topologies suchas, inter alia, buck and boost converters. To improve the switchingefficiency of converters, a drive device is proposed which reducesfreewheeling, i.e. the conduction of the freewheeling diode.Simultaneously, the switching elements are logically locked with respectto each other so as to ensure that they cannot be simultaneouslyconducting, thereby precluding shoot through currents.

In US-A1-2001/0036085 a description is given of a DC/DC converter. Saidconverter is a synchronous buck converter which is “soft-switching”operated, i.e. both switching transitions are “soft”. In this case, thedead time between the switching of the switching elements is controlledwithin the drive circuit. By considering the derivation of the voltageacross the sync-switching element, the dead time is set shorter orlonger, resulting in an “ideal dead time”.

In US-B1-6396250 a description is given of a synchronous buck converterwherein a high efficiency is aimed at. The first and the secondswitching element are driven at a variable dead time. A controlmechanism sets the dead time to a value designated as the optimum value.For the control, the voltage across the second switching element isconsidered. If said voltage exceeds a predetermined threshold valueranging between 0 V and the forward voltage of the freewheeling diode,then a shorter dead time is set by means of a counter.

It is an object of the invention to provide a converter circuit as wellas a drive device and a driving method for same, which enable a stillfurther reduction of the switching losses as compared to the solutionsin accordance with the prior art.

This object is achieved by a converter circuit as claimed in claim 1, adrive device as claimed in claim 9 and a driving method as claimed inclaim 11. Dependent claims relate to advantageous embodiments of theinvention.

The invention is based on the consideration that the losses associatedwith reverse recovery of the freewheeling diode can be precluded orreduced if in the case of a hard switching transition, conduction ofsaid freewheeling diode can be precluded or at least substantiallyprecluded. Said consideration in accordance with the invention impliesthat prior art topologies always had a dead time of positive duration inwhich, upon turn off of the second switching element, the currentpreviously flowing through said second switching element was taken overby the freewheeling diode. If, however, the current is taken over by theother switching element instead, conduction of the freewheeling diodecan be completely precluded under certain conditions. In any case,conduction can be reduced to such an extent that the reverse recoverytime responsible for the recovery current is substantially reduced.

Therefore it is proposed in accordance with the invention that uponswitching from the second switching element to the first switchingelement (i.e. the transition from a first state wherein the secondswitching element conducts and the first switching element does notconduct to a second state wherein the first switching element conductsand the second switching element does not conduct), the timing ofdriving the switching elements is controlled. Said control takes placeby determining whether a shoot through current occurs or thefreewheeling diode conducts. If a shoot through current occurs, thedrive is changed such that the first switching element is turned onlater. If it is determined that the freewheeling diode conducts, thenthe drive is changed such that the first switching element is turned onsooner. The terms “sooner” or “later” are to be interpreted in relationto the turn off of the second switching element. They do not definewhich one of the switching processes (turn on of the first switchingelement, turn off of the second switching element) occurs first. Inprinciple, small dead times are possible (i.e. turn-on of the firstswitching element does not occur until after turn-off of the secondswitching element). Preferably, however, there is a period of overlap(i.e. turn on of the first switching element takes place before turn offof the second switching element).

Consequently, the invention turns away from known solutions that alwayscontained a compulsory dead time between the turn-off of a switchingelement and the turn-on of another switching element. Instead, a realcommutation is proposed wherein the first (control) switch takes overthe load current from the second (sync) switch. The timing is ofdecisive importance here. The control mechanism in accordance with theinvention enables said timing to be set such that ideally shoot throughcurrents and conduction of the freewheeling diode are precluded.

As a result of the fact that conduction of the freewheeling diode isreduced or completely precluded, substantial losses due to reverserecovery do not take place. As a result, a substantial reduction of theswitching losses is achieved, which is important, in particular, duringoperation at high switching frequencies.

The invention can preferably be applied to all converter topologies inwhich the freewheeling path of an inductive element runs via a switchingelement with a parallel freewheeling diode, with the switchingtransition at the switching element being a hard transition. In thisconnection, “hard turn-off” is to be taken to mean that the voltageacross the switching element is changed from the forward to the reversedirection of the freewheeling diode. The freewheeling path of aninductive element is the current path that enables an inductive currentto continue flowing after a switch, whose turn-on has brought about thecurrent built up, has been turned off. The topologies in questioncomprise, for example, a half bridge as well as a fall bridge, as thelatter is composed of two half bridges. Examples of such topologiesinclude, in addition to synchronous buck converters, also synchronousboost converters, synchronous buck-boost converters, synchronous up/downconverters as well as topologies derived therefrom. The switchingelements used in actual circuits will customarily be field effecttransistors wherein the freewheeling diode generally is not a separatecomponent but, as for example with MOSFETs, a property of thesemiconductor switch.

In accordance with a further embodiment of the invention it is providedthat upon switching from the second to the first switching element thetiming is such that a period of overlap is established during which bothswitching elements are simultaneously conducting. In this connection, aswitch embodied so as to be a MOSFET is considered to be conducting ifits gate voltage lies above the threshold voltage. The duration of theperiod of overlap is controlled by determining whether, after turn-offof the second switching element, a shoot through current occurs orconduction of the freewheeling diode occurs. If a shoot through currentoccurs, the duration of the period of overlap is reduced. If thefreewheeling diode becomes conducting, then the duration of the periodof overlap is increased. By virtue of this control strategy, the timingof the drive is optimized.

The voltage across the second switching element can be used as themeasured value for the control. In accordance with a first proposal, thevoltage variation is used to determine whether a shoot through currentor conduction of the freewheeling diode occurs. This is possible, forexample, by determining the absolute minimum value of this voltagewithin a switching interval. This minimum value occurs after turn off ofthe second switching element. If the freewheeling diode is conducting,the voltage drops to its forward voltage for some time. If a shootthrough current occurs, the voltage polarity changes immediately afterturn-off. In a preferred control mechanism, this can be taken intoaccount in a very simple manner in that the timing, for example theperiod of overlap, is set such that the voltage minimum that sets itselfassumes a value between the forward voltage of the switching element andthe forward voltage of the freewheeling diode. The measurement of theminimum voltage is particularly simple because it concerns the detectionof an absolute minimum. To measure such peak values, people skilled inthe art can make use of known means, and the measurement must not belimited to a narrow time range within the switching period.

In accordance with an alternative proposal, the oscillation occurringafter turn off of the second switching element and caused by chargeswitching of the switching capacitance is considered. The amplitude ofthis evanescent oscillation is minimal in the case where neither shootthrough currents nor diode conduction occur. Preferably, the first peakvalue of the oscillation is measured, which is also the absolute maximumof the voltage during the switching period. By controlling the period ofoverlap so that the peak value is minimized, the control aimed at inaccordance with the invention can be realized. Also in this case, thevoltage peak value can be measured using simple means.

When measuring the voltage across a switching element it may occur thatthe accuracy of the measurement outside the housing is influenced, forexample, by housing impedances. Therefore, a further embodiment of aswitching element is proposed in which one or more additional, dedicatedmeasuring lines are provided. While substantial currents flow throughthe available connecting lines, so that for example inductances, such asthey occur at a bonding wire, already have a clear effect, the measuringline only serves to determine a voltage, such as, in the case of aMOSFET, the voltage across the drain-source path. When the voltage ismeasured, the current flowing through the measuring line is so smallthat the measuring result is only negligibly false.

In accordance with a further embodiment of the invention, the timing,i.e. for example the duration of the period of overlap, is controlledsuch that in at least a first switching period a measurement of electricquantities of the converter circuit takes place, on the basis of whichthe duration of the period of overlap is set for a further switchingperiod that is posterior to the first switching period. As a result,also at high frequencies there is sufficient time for setting thetiming. In this connection, it is not necessary for the second switchingperiod to come directly after the first switching period, instead it mayalso be the switching period after next period, or one of the subsequentswitching periods. Preferably, to set the tiring in a switching period,the measured values of a plurality of preceding switching periods areevaluated.

When the converter circuit in accordance with the invention startsoperating, it is preferred that initially a dead time is observedbetween turn-off of the second switching element and turn-on of thefirst element. Since, as discussed above, the correct timing iscritical, it is thus ensured that operation initially begins in anon-critical range, although with initially slightly higher losses. Bythe control in accordance with the invention, the timing of switching ofthe first and the second switching element is changed such that theinitially set dead time is continuously reduced until finally theoptimum is attained at which, if necessary, even a period of overlap isset.

In accordance with a further embodiment of the invention it is providedthat upon switching from the second switching element to the firstswitching element, the first switching element is initially driven suchthat the current flowing through it is limited to a maximum value. Inthe case of a MOSFET this is achieved by driving at a reduced gatevoltage. The maximum current value thus set is above the nominal outputcurrent of the converter circuit. Here it is possible that, for example,values are achieved in a range slightly above the nominal current, forexample approximately 1.2 times the nominal current. The setting of avery high maximum value is also possible, for example a valueapproximately equal to or exceeding 2 times the nominal output current.The maximum value should be selected to be such that the limitation ofthe current thus obtained does not have any effect during normaloperation since this is accompanied by high losses.

These and other aspects of the invention are apparent from and will beelucidated with reference to the embodiment(s) described hereinafter.

IN THE DRAWINGS

FIG. 1 a shows a basic wiring diagram of a synchronous buck converter;

FIG. 1 b shows a basic wiring diagram of a synchronous boost converter;

FIG. 1 c shows a basic switching diagram of a synchronous buck-boostconverter;

FIG. 1 d shows a basic wiring diagram of a synchronous up/downconverter;

FIG. 2 shows a wiring diagram of an embodiment of the buck convertershown in FIG. 1 a;

FIG. 3 shows a schematic diagram to illustrate the variation of currentsand voltages of the circuit of FIG. 2 at a drive with a dead time (inaccordance with the prior art);

FIG. 4 shows a schematic diagram to illustrate the variation of currentsand voltages of the circuit shown in FIG. 2 at a drive with an idealtime of overlap;

FIG. 5 shows the second transition of FIG. 4 on an enlarged scale;

FIG. 6 shows a schematic diagram to illustrate the variation of currentsand voltages of the circuit shown in FIG. 2, in which driving takesplace with too long a period of overlap and in the presence of shootthrough currents;

FIG. 7 shows a schematic diagram illustrating the variation of thevoltage across the second switching element shown in FIG. 2;

FIG. 8 shows a schematic diagram to illustrate the variation of currentsand voltages of the circuit shown in FIG. 1, wherein driving takes placeat a lower gate voltage;

FIG. 9 shows a schematic diagram to illustrate the variation of currentsand voltages of the circuit shown in FIG. 1, wherein driving takes placeat a smaller gate voltage and the shoot through current is limited.

FIGS. 1 a–1 d show converter circuits in accordance with the knowntopologies: buck converter (FIG. 1 a), boost converter (FIG. 1 b),buck-boost converter (FIG. 1 c) and up/down converter (FIG. 1 d). Theconverter circuits 10 each convert an input voltage V_(i) to an outputvoltage V_(o) at the output. The converter circuits 10 each comprise afirst switching element T₁, a second switching element T₂ and aninductive element L. A freewheeling diode D₂ is part of a freewheelingpath for the current through the inductive element L. The switchingelements T₂, which in FIGS. 1 a–1 d are arranged parallel to thefreewheeling diode D₂, serve as synchronous rectifiers, i.e. they aresynchronized with D₂, so that they conduct if the diode D₂ would conductwithout T₂ being present, in order to avoid the losses occurring as aresult of the larger forward voltage.

T₁, T₂, which are ideally represented as switches in FIGS. 1 a–d, inpractice customarily take the form of MOSFETs, wherein the source-drainjunction is switched by applying a gate voltage. In this case,customarily the diodes D₂ are not discrete components but rather theinternal body diodes of the MOSFETs used.

Hereinbelow, an exemplary embodiment of the invention will be explainedin greater detail with regard to the synchronous buck-convertertopology. The synchronous buck converter shown in FIG. 1 a comprises afirst switching element T₁ (control switch) and a second switchingelement T₂ (sync switch) which are connected in the form of a halfbridge 12 to the input voltage V_(i). An inductance L is connected tothe center of the bridge 13, the output V_(o) being situated behind saidinductance. A smoothing capacitor C_(o) is arranged parallel to theoutput. A load (not shown) connected to the output would extend parallelto C_(o). The function of the buck converter of FIG. 1 a in a continuousmode of operation is known to those skilled in the art. The switches T₁,T₂ are driven with voltage pulses, for example a pulse width-modulatedvoltage, so that the output voltage V_(o) obtained is reduced withrespect to the input voltage V_(s), the output voltage beingcontrollable via driving of the switches (for example pulse duty ratio).

FIG. 2 shows a converter circuit 20 being an embodiment of the topologyshown in FIG. 1. T₁, T₂ are embodied so as to be MOSFETs whose gateconnections are each driven by drive circuits 22, 24. A controller 26drives the drivers 22, 24. The controller 26 is also connected to thebridge center 13 and to ground, so that it is capable of detecting thevoltage V_(T2) across the second switching element. The freewheelingdiode D₂ is the internal body diode (drain-source) of the switch T₂, D₂not being separately shown again in FIG. 2. In addition, the switchesT₁, T₂ have parasitic switching capacitances (not shown).

FIG. 3 is a qualitative representation of the variation with respect totime of electrical quantities of the circuit 20 within a switchingperiod T. V_(G1) designates the gate voltage at the first switchingelement T₁, and V_(G2) designates the gate voltage at the secondswitching element T₂. In FIG. 3, the variation with respect to time ofthe currents I_(T1) through the first switching element, I_(T2) throughthe second switching element and I_(D2) through the freewheeling diodeD₂ corresponds only to a qualitative representation that is used to showthe basic variation of these quantities. Variations measured on realcircuits may be different owing to a plurality of parasitic effects.

As shown in FIG. 3, the switches T₁, T₂ are each driven with voltagepulses, for example with a pulse width-modulated voltage. Theirvariation with respect to time (frequency, pulse duty ratio) isestablished by an input signal at control 26, for example to control theoutput voltage V_(o) in known manner. FIG. 3 shows one of the successiveswitching periods T, which switching period first shows a range in whichthe switch T₁ is on, i.e. conducting. In this case, the current I_(L)flows as I_(T1) through the first switching element T₁. The switchingfrom T₁ to T₂ takes place with a first dead time Δt1, in which processT₁ is first turned off and T₂ is not turned until at the end of thefirst dead time Δt1. As the current I_(L) through the inductance Lcannot decrease suddenly, in the first dead time interval Δt1 thefreewheeling diode D₂ becomes conducting at the second switching elementT₂. After turning on T₂, T₂ takes over the current I_(L) as I_(T2), sothat the current I_(D2) through the diode again decreases to zero. Inthe topology shown, this first switching transition from T₁ to T₂ is a“soft” switching transition, wherein turn on of T₂ occurs while itsswitching capacitance is not charged.

In connection with the present invention, the second switching processfrom T₂ to T₁will now be considered. As regards the drive in accordancewith the prior art (FIG. 3), a second dead time Δt2 was always providedfor, in which case first the second switch T₂ is turned off and thefirst switch is not turned on again until after the dead time Δt2 hadended. The second switching transition is a “hard” switching transitionwherein turn on of T₁ occurs at a point in time when its switchingcapacitance is charged to approximately V₁. During the dead time, thecurrent I_(L) is led again through the freewheeling diode D2 after theturn-on of T₁, the diode D2 is operated in the reverse directionhowever. A reverse current (reverse recovery) through the freewheelingdiode D2 then takes place during a reverse recovery time, saidfreewheeling diode conducting in the reverse direction for a shortperiod of time. This reverse current through the diode D2 is designated“RR” in FIG. 3. In this figure however the amplitude as well as theduration of the reverse recovery current are exaggerated for clarity. Asthe variation of the current I_(T1) through the first switching elementshows, the reverse recovery current leads to a clearly symmetricalincrease of I_(T1). This leads to substantial losses in the eachswitching cycle.

FIG. 4 shows a drive in accordance with a first embodiment of theinvention. With this drive the buck converter 20 shown in FIG. 2 isdriven in a special manner at the second switching transition, i.e. uponswitching from the second switching element T₂ to the first switchingelement T₁.

As shown in FIG. 4, when driving takes place in accordance with thefirst embodiment of the invention, no second dead time Δt2 is providedfor. Instead, the half bridge 12 is operated for a short period ofoverlap Δt_(overlap) in such a manner that both T₁ and T₂ areconducting. As shown in FIG. 4, this ideally leads to a commutation ofthe current I_(L) from I_(T2) to I_(T1). If in this case the extent towhich I_(T1) increases is equal to the extent to which I_(T2) decreases,then, in the ideal case shown in FIG. 4, the diode D₂ will not startconducting, so that the current I_(D2) remains zero during the secondswitching transition.

In FIG. 5, the second switching transition of FIG. 4 is shown on anenlarged time scale. The aim of this Figure is to qualitatively depictthe variations of the quantities shown. Said simplified qualitativerepresentation serves to give a better understanding. In theillustrations shown in FIG. 4 and FIG. 5, for example, the chargeswitching of the switching capacitances of the switches T₁, T₂ isignored. In fact, T₁ must carry current for a short, additional periodof time to attain charge switching of the switching capacitances.

The time of overlap Δt_(overlap) comprises the range in which bothswitches T₁, T₂ are simultaneously conducting, i.e. the respective gatevoltages V_(G1), V_(G2) are above the threshold voltages V_(thr) of theMOSFETs. In a concrete embodiment, the time of overlap Δt_(overlap) willbe very short, for example a few nanoseconds.

In FIG. 5, the variation of the gate voltage V_(G1) and V_(G2) upon turnon of the first switching element shows a possible variation of thesevoltages. The actual variation depends on a number of factors (forexample gate-source capacitance, properties of the driver module etc.)and may be different. Likewise, the dependence of the conductance of thedrain-source paths of the two switches on the gate voltages is highlynon-linear. The variation of each of the gate voltages is not decisivehere; what is important is the variation of the currents I_(T2), I_(T1).Ideally, as shown in FIG. 5, a timing can be found that results in anideal commutation from I_(T2) to I_(T1) without the diode D₂ becomingconducting.

With the drive in accordance with the first embodiment of the invention,exact timing is decisive. If the time interval between switching from T₂to T₁ is too long, i.e. if the dead time Δt2 is chosen too long or theperiod of overlap Δt_(overlap) is chosen too short, then the diode D₂will become conducting, as shown in FIG. 3, and subsequently a reverserecovery current with the associated losses will occur. If, on the otherhand, the period of overlap Δt_(overlap) is chosen to be too long, thenthe simultaneous conduction of both switches T₁, T₂ causes a shootthrough current in which current flows as a short-circuit current frominput V_(i) directly through the switches T₁, T₂ (shoot-throughcurrent). This situation is shown in FIG. 6. The long period of overlapΔt_(overlap) chosen here results in a negative current flow I_(T2)through the second switching element T₂. The associated peak also occursin a mirror-inverted manner as a substantially increased current I_(T1)through the first switching element T₁. Such a shoot through currentcauses extremely high losses and may cause damage to the switchingelements T₁, T₂.

Consequently, in a concrete embodiment it is extremely difficult todetermine and set the optimum timing (FIG. 5) in advance because thebehavior depends on many factors, such as the properties of thecomponents, but also on the operating state (load, temperature etc.).Therefore, to obtain the best possible timing to ensure that acommutation from I_(T1) to I_(T2) takes place as shown in FIGS. 4 and 5,the controller 26 is used to control the timing of switching of T₁ andT₂. The controller 26 sets the timing for each switching period T insuch a manner that, on the one hand, conduction of the diode D₂ and thesubsequent reverse-current are precluded, and, on the other hand, also ashoot through current is precluded. Said control provides for a laterturn-on of T₁, i.e. it reduces Δt_(overlap), if a shoot through currentoccurs. If conduction of the diode D₂ is detected upon switching fromthe second switching element T₂ to the first switching element T₁, thenT₁ is turned on sooner, i.e. Δt_(overlap) is increased.

A distinction between, on the one hand, the above-mentioned cases ofdiode conduction, and, on the other hand, shoot through current can bemade by considering the voltage V_(T2) across the second switchingelement T₂. For this purpose, the controller 26 comprises appropriatesinputs. In FIG. 7, the variation of the voltage V_(T2) after turn-off ofthe second switching element T₂ is shown. Three variations with respectto time A, B and C are shown, B being the variation of V_(T2) upon theoccurrence of a shoot through current, C the voltage variation in thecase of diode conduction and A a variation that is aimed at, wherebyboth diode conduction and shoot through currents are precluded. Therepresentation of FIG. 7 is a purely qualitative approach intended toprovide a basis explanation of the interrelationships.

When T₂ is conducting, the voltage V_(T2) has a small, negative valuewhich corresponds to the forward voltage of the switching element T₂,i.e. for example approximately −0.1 V in the case of a MOSFET. Asregards curve C, the diode D₂ starts conducting after turn-off of T₂. Asa result, the voltage T₂ decreases to the forward voltage of the diodeD₂ of, for example, approximately −0.7 V, which is slightly higher thanthe forward voltage of a MOSFET. After the switching element T₁ hascompletely taken over the current I_(L), the diode D₂ is blocked and thevoltage V_(T2) increases. As a result, the switching capacitance of T₂is charged, which leads to the decaying oscillation of V_(T2) shown inFIG. 7 (the switching capacitance of T₂ forms a series-resonant circuitwith continuously available parasitic inductances). Since theoscillation decays, the first maximum {circumflex over (V)}_(C) is themaximum of the voltage. This maximum is too high during conduction ofthe diode D₂. In the absence of a minimum and a maximum of the voltageV_(T2), in the case of curve C, a V_(min, C) is then obtained whichcorresponds to the negative forward voltage of the diode D₂ and arelatively high voltage maximum {circumflex over (V)}_(C) is obtained.

In the case of too long a period of overlap Δt_(overlap) and theresultant shoot through current, the voltage V_(T2) varies approximatelyas shown in curve B. From the negative forward voltage of the switch T₂the voltage rapidly increases without a preceding decrease. Also here adecaying oscillation of the voltage V_(T2) occurs. In this case too, theheight of the first maximum {circumflex over (V)}_(B) depends on thecurrent I_(T2) that has flowed at the moment turn off of T₂ takes place.As this current corresponds to the shoot through current, {circumflexover (V)}_(B) is clearly excessively high when such a current occurs.The curve B is thus characterized by a voltage minimum V_(min, B), whichcorresponds to the forward voltage of T₂, and a high voltage maximum{circumflex over (V)}_(B).

An aimed at variation of V_(T2) that is to be achieved with the aid ofthe control is represented by curve A. From the initially slightlynegative value (forward voltage of T₂), V_(T2) does not rise immediatelyafter turn-off of T₂ (this would be an indication of a shoot throughcurrent), but rather decreases slightly to a value V_(min, A.). Fromthere, V_(T2) rises, while also in this case a decaying oscillationoccurs. The amplitude thereof and hence also the first maximum{circumflex over (V)}_(A) is substantially smaller than in the cases B(shoot through current) and C (diode conduction). Thus, the curve A ischaracterized in respect of minimum and maximum by a small {circumflexover (V)}_(A) and a V_(min, A) that lies between the forward voltage ofT₂ and the forward voltage of the diode D₂.

In the first embodiment of a control 26, the maximum value of thevoltage V_(T2) that adjusts itself after turn off of T₂ is measured. Thecontrol is designed such that the value {circumflex over (V)}_(T2) iscontrolled to a minimum, which could correspond, for example, to curve Ain FIG. 7. The problem encountered with this control is that, at anincreased value of {circumflex over (V)}_(T2), it cannot be readilydetermined whether this can be attributed to too fast a timing (shootthrough current, curve B) or too slow a timing (diode conduction, curveC). This problem can be addressed, however, by approaching the optimumtiming (minimum value of {circumflex over (V)}_(T2)) always from oneside. Thus, as shown in FIG. 2, the timing can initially begin with, forexample, a dead time Δt2. Said dead time is reduced step by step until{circumflex over (V)}_(T2) has reached a minimum value.

In a second, preferred embodiment, the voltage variation of {circumflexover (V)}_(T2) after turn off of T₂ is considered in respect of theminimum that adjusts itself. As explained in connection with FIG. 7, thecurves A, B and C can be clearly distinguished by means of the minimumvalue of V_(T2) obtained after turn off of T₂. The control aims atadjusting the minimum value of V_(T2) to a fixed value V_(min, A) thatlies between the forward voltage of D₂ (V_(min, C)) and the forwardvoltage of T₂ (V_(min, B)) If use is made of a MOSFET, the control couldaim at a predetermined value for V_(min, A) of, for example, −0.3 V. Ifa value of V_(T2min) is obtained that is higher than said value(indication of shoot through current) then the timing is changed suchthat T₁ is turned on later. If a value of V_(T2min) below saidpredetermined value is obtained (indication of diode conduction), thenthe timing is changed such that T₁ is turned on sooner.

The two embodiments of the control 26 shown above are meant as examples.On the one hand, the variation of the voltage V_(T2) as qualitativelyrepresented in FIG. 7 can be observed in a different manner so as todetermine whether shoot through currents or diode conduction occur. Onthe other hand, also other electrical quantities of the circuit 10 canbe detected, such as the current I_(T2), and conclusions regarding thebehavior can be drawn from the data thus obtained. Alternatively, theabove-mentioned criteria for assessing the curve variation of V_(T2) canbe combined to form a sure judgment.

When implementing a concrete control, the controller 26, after havingbeen turned on, initially operates such that a drive operation with adead time (FIG. 3) takes place. In each switching period T, electricalquantities of the circuit 20 are then observed as indicated hereinabove.On the basis of said observations in one or more switching periods, thespecified value for the timing in the subsequent (or one of thesubsequent) switching periods is set by means of the above-mentionedcontrol. Starting out from an initially substantial dead time after turnon, this leads to a reduction of said dead time until the desired resultis attained, i.e. an optimum timing with direct commutation from T₂ toT₁. This result of the control will probably be achieved at a dead-timerange of negative duration, i.e. a short period of overlap Δt_(overlap).The timing is controlled more and more, so that a change in theoperating conditions, for example a load change, triggers a rapidreaction.

In a third embodiment of the invention, the first switching element T₁is driven in a protection period after turn on at a reduced gatevoltage. By driving at a suitably reduced gate voltage, the currentflowing through a MOSFET can be limited to a maximum value. However, ifthis limitation becomes active, i.e. if, without the low gate voltage, ahigher current would flow than that corresponding to the maximum value,then, at the MOSFET, an increased voltage drop and a correspondinglyhigh power dissipation occur. Therefore, for the third embodiment of theinvention, it is proposed to drive at a gate voltage such that theresultant maximum current through the first switching element T₁ ishigher than the current I_(Nenn) flowing through T_(i) during nominaloperation. Thus, this type of drive serves to limit increased currentsI_(T1) that occur in connection with shoot through currents.

A corresponding drive is qualitatively shown in FIG. 8. For a shortperiod of time Δtp, which includes the turn-on instant of T₁, the gatevoltage V_(G1) at the first switching element T₁ is not set to themaximum value but only to a reduced value V_(G1, p). As this valueV_(G1, p) is chosen to be so high that the current I_(T2), which canmaximally reach the nominal current I_(Nenn) and hence is below thethreshold value I_(T1, p) specified thereby, is not influenced, thechanged drive does not have an effect on the normal operation depictedin FIG. 8.

If, however, a longer period of overlap Δt_(overlap), as shown in FIG.9, causes a substantial shoot through current, then this shoot throughcurrent is limited to the maximum value I_(T1, p) specified by thereduced gate voltage U_(G1,p). As a result, substantial losses in T₁occur. However, the risk of destructions caused by correspondingexcessive currents no longer exists.

The value for I_(T1, p) is specified such that the limitation becomesactive as rarely as possible. I_(T1, p) can be set to, for example, 2times the nominal current I_(Nenn) at the output of the convertercircuit. In this case, the drive in accordance with the third embodimentserves as a protective mechanism which in the event that theabove-mentioned control cannot effectively preclude an excessive currentdue to, for example, load changes or other effects, protects the circuitagainst destruction.

It is also possible, however, that a lower value for I_(T1,p) is set,for example approximately 1.2 to 1.5 times the nominal current I_(Nenn).This makes it possible, in addition to protection against damage byexcessive currents, for example to reduce the amplitude of theoscillation of V_(T2) and hence the emission of electromagneticinterference.

The above-described embodiments of the invention are explained withreference to synchronous buck converters. The mode of driving, thecontrol methods and the current limitation can also be applied, however,in any desired combination in other converter topologies (FIGS. 1 b–1d). In all topologies, the controller 26 that sets the drive of theswitches can, on the one hand, control the output voltage in a knownmanner and, on the other hand, set the timing of switching such that theswitching losses are minimized.

1. A converter circuit comprising: at least a first switching clement(T₁) and a second switching element (T₂) and an inductive element (L),wherein a control device (26) is provided to alternately switch theswitching elements (T₁, T₂) so that a current (I_(L)) flows through theinductive element (L), and wherein at least at the second switchingelement (T₂) there is provided a freewheeling diode (D₂) which iscapable of conducting the current flowing through the inductive element(L) after turn-off of the first switching element (T₁), wherein thecontrol device (26) controls a timing of driving the switching elements(T₁, T₂) upon switching front the second switching element (T₂) to thefirst switching element (T₁) by determining whether one of a shootthrough current occurs and the freewheeling diode (D₂) is conducting,wherein, upon the occurrence of the shoot through current, the timing ofdriving the switching elements (T₁, T₂) is changed such that the turn onof the first switching element (T₁) takes place later with respect tothe instant of turn off of the second switching element (T₂), and whenthe freewheeling diode (D₂) is conducting, the timing of driving theswitching elements (T₁, T₂) is changed such that the turn on of thefirst switching element (T₁) takes place sooner with respect to theinstant of turn off of the second switching element (T₂).
 2. A convertercircuit as claimed in claim 1, wherein the switching elements (T₁, T₂)are driven such that they are simultaneously conducting during a periodof overlap (Δ_(Toverlap)), and wherein the control device (26) controlsthe duration of the period of overlap (Δ_(Toverlap)) in that it isdetermined whether one of the shoot through current occurs and thefreewheeling diode (D₂) is conducting, wherein, upon the occurrence ofthe shoot through current, the duration of the period of overlap isreduced, and, when the freewheeling diode (D₂) is conducting, theduration of the period of overlap is increased.
 3. A converter circuitas claimed in claim 1, wherein the control device (26) comprises meansfor measuring a voltage (V_(T2)) across the second switching element(T₂), the voltage (V_(T2)) being observed at least after turn-off of thesecond switching element (T₂), and it is determined, by means of avoltage variation, whether one of the shoot through current occurs andthe freewheeling diode (D₂) is conducting.
 4. A converter circuit asclaimed in claim 3, wherein the second switching element (T₂) is aMOSEET in a housing. wherein at least connecting lines for a drain, asource and a gate of the MOSFET are led from the housing to an exterior,wherein one or more measuring lines are provided for determining thevoltage (V_(T2)) between the drain and the source.
 5. A convener circuitas claimed in claim 3, wherein a peak value ({circumflex over (V)}_(r2))is determined from an oscillating voltage obtained after turn-off of thesecond switching element (T₂). and the timing of the drive of theswitching elements (T₁, T₂) is set such that said peak value({circumflex over (V)}_(r2)) is minimized.
 6. A converter circuit asclaimed in claim 3, wherein a minimum value of the voltage (V_(T2))across the second switching element (T₂) is determined, and the timingof driving the switching elements (T₁, T₂) is set such that the minimumvalue of the voltage lies between a forward voltage of the secondswitching element (T₂) and a forward voltage of the freewheeling diode(D₂).
 7. A converter circuit as claimed in claim 1, wherein the controldevice (26) comprises means for measuring at least one electricalquantity (V_(T2)) of the converter circuit (12), in the course of atleast a first switching period (T) at least one measurement is carriedout, and said measurement is sued to set the timing of driving theswitching elements (T₁, T₂) in a second switching period.
 8. A convertercircuit as claimed in claim 1, wherein at an onset of operation, uponswitching from the second to the first switching element, a dead time isprovided between the turn off of the second switching element (T₂) andthe turn on of the firs switching element (T₁).
 9. A converter circuitas claimed in claim 1, wherein upon switching from the second switchingelement (T₂) to the first switching element (T₁) the first switchingelement (T₁) is driven in such a way, for a protection period that lastsat least until the turn off of the second switching element (T₂), thatthe current through the first switching element (T₁) cannot exceed athreshold value (I_(t1, max)), which threshold value (I_(t1, max)) liesabove a nominal output current of the converter circuit.
 10. A drivedevice for alternatively switching a first switching element (T₁) and asecond switching element (T₂) so that a current (I_(t)) flows through aninductive element (L), the second switching element (T₂) being providedwith a freewheeling diode (D₂) which is capable of conducting thecurrent (I₁) flowing through the inductive element (L) after turn-off ofthe first switching element (T₁), the drive device comprising: a pair ofdrive circuits (24, 25) for driving the first and second switchingelements (T₁, T₂), and a control device (26) for determining whether oneof a shoot through current occurs and a freewheeling diode (D₂) isconducting, the control device (26) controls the drive circuits (24, 25)wherein a timing of driving the first and second switching elements (T₁,T₂) upon switching from the second switching element (T₂) to the firstswitching element (T₁) is controlled such that, upon the occurrence ofthe shoot through current, the timing of driving the switching elements(T₁, T₂) is changed such that the turn on of the first switching element(T₁) takes place later than the instant of turn off of the secondswitching element (T₂), and when the freewheeling diode (D₂) isconducting, the timing of driving the switching elements (T₁, T₂) ischanged such that the turn on of the first switching element (T₁) takesplace before the instant of turn off of the second switching element(T₂).
 11. A drive method for a convener switch comprising at least onehalf bridge (12) with a first and a second switching element (T₁, T₂),in which at least at the second switching element (T₂) and afreewheeling diode (D₂) is provided, wherein a timing of switching ofthe switching elements (T₁, T₂) upon switching from the second switchingelement (T₂) to the first switching element (T₁) is controlled, whereinit is determined whether one of the freewheeling diode (D₂) conducts anda shoot through current occurs, wherein, upon the occurrence of theshoot through current, the turn on of the first switching element (T₁)rakes place later with respect to the instant of turn off of the secondswitching element (T₂), and when the freewheeling diode (D₂) isconducting, the turn on of the first switching element (T₁) takes placesooner with respect to the instant of turn off of the second switchingelement (T₂).